Method and apparatus for processing data for transmission in a multi-channel communication system using selective channel inversion

ABSTRACT

Techniques to process data for transmission over a set of transmission channels selected from among all available transmission channels. In an aspect, the data processing includes coding data based on a common coding and modulation scheme to provide modulation symbols and pre-weighting the modulation symbols for each selected channel based on the channel&#39;s characteristics. The pre-weighting may be achieved by “inverting” the selected channels so that the received SNRs are approximately similar for all selected channels. With selective channel inversion, only channels having SNRs at or above a particular threshold are selected, “bad” channels are not used, and the total available transmit power is distributed across only “good” channels. Improved performance is achieved due to the combined benefits of using only the NS best channels and matching the received SNR of each selected channel to the SNR required by the selected coding and modulation scheme.

PRIORITY CLAIM

This application is a continuation application of, and claims thebenefit of priority from, U.S. patent application Ser. No. 09/860,274,entitled “Method and Apparatus for Processing Data for Transmission in aMulti-Channel Communication System Using Selective” and filed on May 17,2001, which is fully incorporated herein by reference for all purposes.

BACKGROUND

1. Field

The present invention relates generally to data communication, and morespecifically to a novel and improved method and apparatus for processingdata for transmission in a wireless communication system using selectivechannel inversion.

2. Background

A multi-channel communication system is often deployed to provideincreased transmission capacity for various types of communication suchas voice, data, and so on. Such a multi-channel system may be amultiple-input multiple-output (MIMO) communication system, anorthogonal frequency division modulation (OFDM) system, a MIMO systemthat utilizes OFDM, or some other type of system. A MIMO system employsmultiple transmit antennas and multiple receive antennas to exploitspatial diversity to support a number of spatial subchannels, each ofwhich may be used to transmit data. An OFDM system effectivelypartitions the operating frequency band into a number of frequencysubchannels (or frequency bins), each of which is associated with arespective subcarrier on which data may be modulated. A multi-channelcommunication system thus supports a number of “transmission” channels,each of which may correspond to a spatial subchannel in a MIMO system, afrequency subchannel in an OFDM system, or a spatial subchannel of afrequency subchannel in a MIMO system that utilizes OFDM.

The transmission channels of a multi-channel communication systemtypically experience different link conditions (e.g., due to differentfading and multipath effects) and may achieve differentsignal-to-noise-plus-interference ratios (SNRs). Consequently, thetransmission capacities (i.e., the information bit rates) that may besupported by the transmission channels for a particular level ofperformance may be different from channel to channel. Moreover, the linkconditions typically vary over time. As a result, the bit ratessupported by the transmission channels also vary with time.

The different transmission capacities of the transmission channels plusthe time-variant nature of these capacities make it challenging toprovide an effective coding and modulation scheme capable of processingdata prior to transmission on the channels. Moreover, for practicalconsiderations, the coding and modulation scheme should be simple toimplement and utilize at both the transmitter and receiver systems.

There is therefore a need in the art for techniques to effectively andefficiently process data for transmission on multiple transmissionchannels with different capacities.

SUMMARY

Aspects of the invention provide techniques to process data fortransmission over a set of transmission channels selected from among allavailable transmission channels. In an aspect, the data processingincludes coding data based on a common coding and modulation scheme toprovide modulation symbols and pre-weighting the modulation symbols foreach selected transmission channel based on the channel'scharacteristics. The pre-weighting may be achieved by “inverting” theselected transmission channels so that thesignal-to-noise-plus-interference ratios (SNRs) are approximatelysimilar at a receiver system for all selected transmission channels. Inone embodiment, which is referred to as selective channel inversion(SCI), only transmission channels having SNRs (or power gains) at orabove a particular SNR (or power gain) threshold are selected for datatransmission, and “bad” transmission channels are not used. Withselective channel inversion, the total available transmit power isdistributed (unevenly) across only “good” transmission channels, andimproved efficiency and performance are achieved. In another embodiment,all available transmission channels are selected for use and the channelinversion is performed for all available channels.

The channel inversion techniques simplify the coding/modulation at thetransmitter system and the decoding/demodulation at the receiver system.Moreover, the selective channel inversion technique may also provideimproved performance due to the combined benefits of (1) using only theN_(S) best transmission channels selected from among all availabletransmission channels and (2) matching the received SNR of each selectedtransmission channel to the SNR required by the coding and modulationscheme selected for use.

The invention further provides methods, systems, and apparatus thatimplement various aspects, embodiments, and features of the invention,as described in further detail below.

BRIEF DESCRIPTION OF THE DRAWINGS

The features, nature, and advantages of the present invention willbecome more apparent from the detailed description set forth below whentaken in conjunction with the drawings in which like referencecharacters identify correspondingly throughout and wherein:

FIG. 1 is a diagram of a multiple-input multiple-output (MIMO)communication system that may be designed and operated to implementvarious aspects and embodiments of the invention;

FIG. 2A is a flow diagram of a process to determine the amount oftransmit power to be allocated to each selected transmission channelbased on selective channel inversion, in accordance with an embodimentof the invention;

FIG. 2B is a flow diagram of a process to determine a threshold a usedto select transmission channels for data transmission, in accordancewith an embodiment of the invention;

FIG. 3 is a diagram of a MIMO communication system capable ofimplementing various aspects and embodiments of the invention;

FIGS. 4A, 4B, and 4C are block diagrams of three MIMO transmittersystems capable of processing data in accordance with three specificembodiments of the invention;

FIG. 5 is a block diagrams of a MIMO receiver system capable ofreceiving data in accordance with an embodiment of the invention;

FIGS. 6A and 6B are block diagrams of an embodiment of a channelMIMO/data processor and an interference canceller, respectively, withinthe MIMO receiver system shown in FIG. 5; and

FIG. 7 is a block diagrams of a MIMO receiver system capable ofreceiving data in accordance with another embodiment of the invention.

DETAILED DESCRIPTION

Various aspects, embodiments, and features of the invention may beapplied to any multi-channel communication system in which multipletransmission channels are available for data transmission. Suchmulti-channel communication systems include multiple-inputmultiple-output (MIMO) systems, orthogonal frequency division modulation(OFDM) systems, MIMO systems that utilize OFDM, and others. Themulti-channel communication systems may also implement code divisionmultiple access (CDMA), time division multiple access (TDMA), frequencydivision multiple access (FDMA), or some other multiple accesstechniques. Multiple access communication systems can support concurrentcommunication with a number of terminals (i.e., users).

FIG. 1 is a diagram of a multiple-input multiple-output (MIMO)communication system 100 that may be designed and operated to implementvarious aspects and embodiments of the invention. MIMO system 100employs multiple (N_(T)) transmit antennas and multiple (N_(R)) receiveantennas for data transmission. MIMO system 100 is effectively formedfor a multiple access communication system having a base station (BS)104 that concurrently communicates with a number of terminals (T) 106.In this case, base station 104 employs multiple antennas and representsthe multiple-input (MI) for uplink transmissions and the multiple-output(MO) for downlink transmissions. The downlink (i.e., forward link)refers to transmissions from the base station to the terminals, and theuplink (i.e., reverse link) refers to transmissions from the terminalsto the base station.

A MIMO system employs multiple (N_(T)) transmit antennas and multiple(N_(R)) receive antennas for data transmission. A MIMO channel formed bythe N_(T) transmit and N_(R) receive antennas may be decomposed intoN_(C) independent channels, with N_(C)≦min {N_(T), N_(R)}. Each of theN_(C) independent channels is also referred to as a spatial subchannelof the MIMO channel and corresponds to a dimension. In one common MIMOsystem implementation, the N_(T) transmit antennas are located at andassociated with a single transmitter system, and the N_(R) receiveantennas are similarly located at and associated with a single receiversystem. A MIMO system may also be effectively formed for a multipleaccess communication system having a base station that concurrentlycommunicates with a number of terminals. In this case, the base stationis equipped with a number of antennas and each terminal may be equippedwith one or more antennas.

An OFDM system effectively partitions the operating frequency band intoa number of (N_(F)) frequency subchannels (i.e., frequency bins). Ateach time slot, a modulation symbol may be transmitted on each of theN_(F) frequency subchannels. Each time slot corresponds to a particulartime interval that may be dependent on the bandwidth of the frequencysubchannel.

A multi-channel communication system may be operated to transmit datavia a number of transmission channels. For a MIMO system not utilizingOFDM, there is typically only one frequency subchannel and each spatialsubchannel may be referred to as a transmission channel. For a MIMOsystem utilizing OFDM, each spatial subchannel of each frequencysubchannel may be referred to as a transmission channel. And for an OFDMsystem not utilizing MIMO, there is only one spatial subchannel for eachfrequency subchannel and each frequency subchannel may be referred to asa transmission channel.

The transmission channels in a multi-channel communication systemtypically experience different link conditions (e.g., due to differentfading and multipath effects) and may achieve differentsignal-to-noise-plus-interference ratios (SNRs). Consequently, thecapacity of the transmission channels may be different from channel tochannel. This capacity may be quantified by the information bit rate(i.e., the number of information bits per modulation symbol) that may betransmitted on a transmission channel for a particular level ofperformance (e.g., a particular bit error rate (BER) or packet errorrate (PER)). Since the link conditions typically vary with time, thesupported information bit rates for the transmission channels also varywith time.

To more fully utilize the capacity of the transmission channels, channelstate information (CSI) descriptive of the link conditions may bedetermined (typically at the receiver system) and provided to thetransmitter system. The transmitter system may then process (e.g.,encode, modulate, and pre-weight) data such that the transmittedinformation bit rate for each channel matches the transmission capacityof the channel. CSI may be categorized as either “full CSI” or “partialCSI”. Full CSI includes sufficient characterization (e.g., the amplitudeand phase) across the entire system bandwidth for the propagation pathbetween each transmit-receive antenna pair in a N_(T)×N_(R) MIMO matrix(i.e., the characterization for each transmission channel). Partial CSImay include, for example, the SNRs of the transmission channels.

Various techniques may be used to process data prior to transmissionover multiple transmission channels. In one technique, data for eachtransmission channel may be coded and modulated based on a particularcoding and modulation scheme selected for that channel based on thechannel's CSI. By coding and modulating separately for each transmissionchannel, the coding and modulation may be optimized for the SNR achievedby each channel. In one implementation of such a technique, a fixed basecode is used to encode data, and the coded bits for each transmissionchannel are then punctured (i.e., selectively deleted) to obtain a coderate supported by that channel. In this implementation, the modulationscheme for each transmission channel is also selected based on thechannel's code rate and SNR. This coding and modulation implementationis described in further detail in U.S. Pat. No. 6,961,388, entitled“CODING SCHEME FOR A WIRELESS COMMUNICATION SYSTEM,” filed Feb. 1, 2001,assigned to the assignee of the present application and incorporatedherein by reference. For this first technique, substantialimplementation complexity is typically associated with having adifferent code rate and modulation scheme for each transmission channel.

In accordance with an aspect of the invention, techniques are providedto (1) process data for all selected transmission channels based on acommon coding and modulation scheme to provide modulation symbols, and(2) pre-weight the modulation symbols for each selected transmissionchannel based on the channel's CSI. The pre-weighting may be achieved byinverting the selected transmission channels so that, in general, theSNRs are approximately similar at the receiver system for all selectedtransmission channels. In one embodiment, which is referred to asselective channel inversion (SCI), only transmission channels havingSNRs (or power gains) at or above a particular SNR (or power gain)threshold are selected for data transmission, and “bad” transmissionchannels are not used. With selective channel inversion, the totalavailable transmit power is distributed across only “good” transmissionchannels, and improved efficiency and performance are achieved. Inanother embodiment, all available transmission channels are selected foruse and the channel inversion is performed for all available channels.

These channel inversion techniques may be advantageously used when fullor partial CSI is available at the transmitter. These techniquesameliorate most of the complexity associated with the channel-specificcoding and modulation technique described above, while still achievinghigh performance. Moreover, the selective channel inversion techniquemay also provide improved performance over the channel-specific codingand modulation technique due to the combined benefits of (1) using onlythe N_(S) best transmission channels from among the availabletransmission channels and (2) matching the received SNR of each selectedtransmission channel to the SNR required for the selected coding andmodulation scheme.

For a MIMO system utilizing OFDM and having full-CSI available, thetransmitter system has knowledge of the complex-valued gain of thetransmission path between each transmit-receive antenna pair of eachfrequency subchannel. This information may be used to render the MIMOchannel orthogonal so that each eigen-mode (i.e., spatial subchannel)may be used for an independent data stream.

For a MIMO system utilizing OFDM and having partial-CSI available, thetransmitter has limited knowledge of the transmission channels.Independent data streams may be transmitted on correspondingtransmission channels over the available transmit antennas, and thereceiver system uses a particular linear or non-linear processingtechnique (i.e., equalization) to separate out the data streams. Theequalization provides an independent data stream corresponding to eachtransmission channel (i.e., each transmit antenna and/or each frequencysubchannel), and each of these data streams has an associated SNR.

If the set of SNRs for the transmission channels is available at thetransmitter system, this information may be used to select the propercoding and modulation scheme and distribute the total available transmitpower. In an embodiment, the available transmission channels are rankedin order of decreasing SNR, and the total available transmit power isallocated to and used for the N_(S) best transmission channels. In anembodiment, transmission channels having SNRs that fall below aparticular SNR threshold are not selected for use. The SNR threshold maybe selected to optimize throughput or some other criteria. The totalavailable transmit power is distributed across all transmission channelsselected for use so that the transmitted data streams have approximatelysimilar SNRs at the receiver system. Similar processing may be performedif the channel gains are available at the transmitter system. In anembodiment, a common coding scheme (e.g., a particular Turbo code of aparticular code rate) and a common modulation scheme (e.g., a particularQAM constellation) are used for all selected transmission channels.

Transmission Channel Inversion

If a simple (common) coding and modulation scheme can be used at thetransmitter system, a single (e.g., convolutional or Turbo) coder andcode rate may be used to encode data for all transmission channelsselected for data transmission, and the resultant coded bits may bemapped to modulation symbols using a single (e.g., PSK or QAM)modulation scheme. The resultant modulation symbols are then all drawnfrom the same “alphabet” of possible modulation symbols and encoded withthe same code and code rate. However, the transmission channels in amulti-channel communication system typically experience different linkconditions and achieve different SNRs. In this case, if the same amountof transmit power is used for each selected transmission channel, thetransmitted modulation symbols will be received at different SNRsdepending on the specific channels on which the modulation symbols aretransmitted. The result will be a large variation in symbol errorprobability over the set of selected transmission channels, and anassociated loss in bandwidth efficiency.

In accordance with an aspect of the invention, a power control mechanismis used to set or adjust the transmit power level for each transmissionchannel selected for data transmission to achieve a particular SNR atthe receiver system. By achieving similar received SNRs for all selectedtransmission channels, a single coding and modulation scheme may be usedfor all selected transmission channels, which can greatly reduce thecomplexity of the coding/modulation process at the transmitter systemand the demodulation/decoding process at the receiver system. The powercontrol may be achieved by “inverting” the selected transmissionchannels and properly distributing the total available transmit poweracross all selected channels, as described in further detail below.

If the same amount of transmit power is used for all availabletransmission channels, then the received power for a particular channelmay be expressed as:

$\begin{matrix}{{P_{rx}^{\prime}\left( {j,k} \right)} = {\frac{P_{tx}}{N_{T}N_{F}}{{H\left( {j,k} \right)}}^{2}}} & {{Eq}\mspace{14mu} (1)}\end{matrix}$

where P_(rx)′(j,k) is the received power for transmission channel (j,k)(i.e., the j-th spatial subchannel of the k-th frequency subchannel),P_(tx) is the total transmit power available at the transmitter, N_(T)is the number of transmit antennas, N_(F) is the number of frequencysubchannels, and H(j,k) is the complex-valued “effective” channel gainfrom the transmitter system to the receiver system for transmissionchannel (j,k). For simplicity, the channel gain H(j,k) includes theeffects of the processing at the transmitter and receiver. Also forsimplicity, it is assumed that the number of spatial subchannels isequal to the number of transmit antennas and N_(T)·N_(F) represents thetotal number of available transmission channels. If the same amount ofpower is transmitted for each available transmission channel, the totalreceived power P_(rx) for all available transmission channels may beexpressed as:

$\begin{matrix}{P_{rx} = {\sum\limits_{j = 1}^{N_{T}}{\sum\limits_{k = 1}^{N_{F}}{\frac{P_{tx}}{N_{T}N_{F}}{{{H\left( {j,k} \right)}}^{2}.}}}}} & {{Eq}\mspace{14mu} (2)}\end{matrix}$

Equation (1) shows that the receive power for each transmission channelis dependent on the power gain of that channel, i.e., |H(j,k)|². Toachieve equal received power across all available transmission channels,the modulation symbols for each channel can be pre-weighted at thetransmitter by a weight of W(j,k), which can be expressed as:

$\begin{matrix}{{{W\left( {j,k} \right)} = \frac{c}{{H\left( {j,k} \right)}}},} & {{Eq}\mspace{14mu} (3)}\end{matrix}$

where c is a factor chosen such that the received powers for alltransmission channels are approximately equal at the receiver. As shownin equation (3), the weight for each transmission channel is inverselyproportional to that channel's gain. The weighted transmit power fortransmission channel (j,k) can then be expressed as:

$\begin{matrix}{{{P_{tx}\left( {j,k} \right)} = \frac{{bP}_{tx}}{{{H\left( {j,k} \right)}}^{2}}},} & {{Eq}\mspace{14mu} (4)}\end{matrix}$

where b is a “normalization” factor used to distribute the totaltransmit power among the available transmission channels. Thisnormalization factor b can be expressed as:

$\begin{matrix}{{b = \frac{1}{\sum\limits_{j = 1}^{N_{T}}{\sum\limits_{k = 1}^{N_{F}}{{H\left( {j,k} \right)}}^{- 2}}}},} & {{Eq}\mspace{14mu} (5)}\end{matrix}$

where c²=b. As shown in equation (5), the normalization factor b iscomputed as the sum of the reciprocal power gain for all availabletransmission channels.

The pre-weighting of the modulation symbols for each transmissionchannel by W(j,k) effectively “inverts” the transmission channel. Thischannel inversion results in the amount of transmit power for eachtransmission channel being inversely proportional to the channel's powergain, as shown in equation (4), which then provides a particularreceived power at the receiver. The total transmit power is thuseffectively distribute (unevenly) to all available transmission channelsbased on their channel gains such that all transmission channels haveapproximately equal received power, which may be expressed as:

P _(rx)(j,k)=bP _(tx).  Eq (6)

If the noise variance is the same across all transmission channels, thenthe equal received power allows the modulation symbols for all channelsto be generated based on a single common coding and modulation scheme,which then greatly simplify the coding and decoding processes.

If all available transmission channels are used for data transmissionregardless of their channel gains, then the poor transmission channelsare allocated more of the total transmit power. In fact, to achievesimilar received power for all transmission channels, the poorer atransmission channel gets the more transmit power needs to be allocatedto this channel. When one or more transmission channels get too poor,the amount of transmit power needed for these channels would deprive (orstarve) the good channels of power, which may then dramatically decreasethe overall system throughput.

Selective Channel Inversion Based on Channel Gains

In an aspect, the channel inversion is applied selectively, and onlytransmission channels whose received power is at or above a particularthreshold, α, relative to the total received power are selected for datatransmission. Transmission channels whose received power falls belowthis threshold are erased (i.e., not used). For each selectedtransmission channel, the modulation symbols are pre-weighted at thetransmitter such that all selected transmission channels are received atapproximately similar power level. The threshold can be selected tomaximize throughput or based on some other criteria. The selectivechannel inversion scheme preserves most of the simplicity inherent inusing a common coding and modulation scheme for all transmissionchannels while also providing high performance associated withseparately coding per transmission channel.

Initially, the average power gain, L_(ave), is computed for allavailable transmission channels and can be expressed as:

$\begin{matrix}{L_{ave} = {\frac{\sum\limits_{j = 1}^{N_{T}}{\sum\limits_{k = 1}^{N_{F}}{{H\left( {j,k} \right)}}^{2}}}{N_{T}N_{F}}.}} & {{Eq}\mspace{14mu} (7)}\end{matrix}$

The modulation symbols for each selected transmission channel can bepre-weighted at the transmitter by a weight of {tilde over (W)}(j,k),which can be expressed as:

$\begin{matrix}{{\overset{\sim}{W}\left( {j,k} \right)} = {\frac{\overset{\sim}{c}}{{H\left( {j,k} \right)}}.}} & {{Eq}\mspace{14mu} (8)}\end{matrix}$

The weight for each selected transmission channel is inverselyproportional to that channel's gain and is determined such that allselected transmission channels are received at approximately equalpower. The weighted transmit power for each transmission channel canthen be expressed as:

$\begin{matrix}{{P_{tx}\left( {j,k} \right)} = \left\{ \begin{matrix}{\frac{\overset{\sim}{b}P_{tx}}{{{H\left( {j,k} \right)}}^{2}},} & {{{H\left( {j,k} \right)}}^{2} \geq {\alpha \; L_{ave}}} \\{0,} & {{otherwise},}\end{matrix} \right.} & {{Eq}\mspace{14mu} (9)}\end{matrix}$

where α is the threshold and {tilde over (b)} is a normalization factorused to distribute the total transmit power among the selectedtransmission channels. As shown in equation (9), a transmission channelis selected for use if its power gain is greater than or equal to apower gain threshold (i.e., |H(j,k)|²≧αL_(ave)). The normalizationfactor {tilde over (b)} is computed based on only the selectedtransmission channels and can be expressed as:

$\begin{matrix}{\overset{\sim}{b} = {\frac{1}{\sum\limits_{{{H{({j,k})}}}^{2} \geq {\alpha \; L_{ave}}}{{H\left( {j,k} \right)}}^{- 2}}.}} & {{Eq}\mspace{14mu} (10)}\end{matrix}$

Equations (7) through (10) effectively distribute the total transmitpower to the selected transmission channels based on their power gainssuch that all selected transmission channels have approximately equalreceived power, which may be expressed as:

$\begin{matrix}{{P_{rx}\left( {j,k} \right)} = \left\{ \begin{matrix}{{\overset{\sim}{b}P_{tx}},} & {{{H\left( {j,k} \right)}}^{2} \geq {\alpha \; L_{ave}}} \\{0,} & {{otherwise},}\end{matrix} \right.} & {{Eq}\mspace{14mu} (11)}\end{matrix}$

Selective Channel Inversion Based on Channel SNRs

In many systems, the known quantities at the receiver system are thereceived SNRs for the transmission channels rather than the channelgains (i.e., the path losses). In such systems, the selective channelinversion technique can be readily modified to operate based on thereceived SNRs instead of the channel gains.

If equal transmit power is used for all available transmission channelsand the noise variance, σ², is constant for all channels, then thereceived SNR for transmission channel (j,k) can be expressed as:

$\begin{matrix}{{\gamma \left( {j,k} \right)} = {\frac{P_{rx}\left( {j,k} \right)}{\sigma^{2}} = {\frac{P_{tx}}{\sigma^{2}N_{T}N_{F}}{{{H\left( {j,k} \right)}}^{2}.}}}} & {{Eq}\mspace{14mu} (12)}\end{matrix}$

The average received SNR, γ_(ave), for each available transmissionchannel may be expressed as:

$\begin{matrix}{{\gamma_{ave} = {\frac{P_{tx}}{{\sigma^{2}\left( {N_{T}N_{F}} \right)}^{2}}{\sum\limits_{j = 1}^{N_{T}}{\sum\limits_{k = 1}^{N_{F}}{{H\left( \mspace{11mu} {j,k} \right)}}^{2}}}}},} & {{Eq}\mspace{14mu} (13)}\end{matrix}$

which also assumes equal transmit power over the available transmissionchannels. The received SNR, S, for all available transmission channelsmay be expressed as:

$\begin{matrix}{S = {{\frac{P_{tx}}{\sigma^{2}}L_{ave}} = {\frac{P_{tx}}{\sigma^{2}N_{T}N_{F}}{\sum\limits_{j = 1}^{N_{T}}{\sum\limits_{k = 1}^{N_{F}}{{{H\left( \mspace{11mu} {j,k} \right)}}^{2}.}}}}}} & {{Eq}\mspace{14mu} (14)}\end{matrix}$

The received SNR, S, is based on the total transmit power being equallydistributed across all available transmission channels.

A normalization factor, β, used to distribute the total transmit poweramong the selected transmission channels can be expressed as:

$\begin{matrix}{\beta = {\frac{1}{\sum\limits_{{\gamma {({j,k})}} \geq {\alpha\gamma}_{ave}}{\gamma \left( {j,k} \right)}^{- 1}}.}} & {{Eq}\mspace{14mu} (15)}\end{matrix}$

As shown in equation (15), the normalization factor β is computed basedon, and as the sum of the reciprocal of, the SNRs of all selectedtransmission channels.

To achieve similar received SNR for all selected transmission channels,the modulation symbols for each selected transmission channel (j,k) maybe pre-weighted by a weight that is related to that channel's SNR, whichmay be expressed as:

$\begin{matrix}{{\overset{\sim}{W}\left( {j,k} \right)} = {\frac{\overset{\sim}{c}}{\sqrt{\gamma \left( {j,k} \right)}}.}} & {{Eq}\mspace{14mu} (16)}\end{matrix}$

where {tilde over (c)}²=β. The weighted transmit power for eachtransmission channel may then be expressed as:

$\begin{matrix}{{P_{tx}\left( {j,k} \right)} = \left\{ \begin{matrix}{\frac{\beta \; P_{tx}}{\gamma \left( {j,k} \right)},} & {{\gamma \left( {j,k} \right)} \geq {\alpha \; \gamma_{ave}}} \\{0,} & {{otherwise}.}\end{matrix} \right.} & {{Eq}\mspace{14mu} (17)}\end{matrix}$

As shown in equation (17), only transmission channels for which thereceived SNR is greater than or equal to an SNR threshold (i.e.,γ(j,k)≧αγ_(ave)) is selected for use.

If the total transmit power is distributed across all selectedtransmission channels such that the receive SNR is approximately similarfor all selected channels, then the resulting received SNR for eachtransmission channel may be expressed as:

$\begin{matrix}{{\overset{\sim}{\gamma}\left( {j,k} \right)} = \left\{ \begin{matrix}{\frac{\beta \; S}{\gamma_{ave}},} & {{\gamma \left( {j,k} \right)} \geq {\alpha \; \gamma_{ave}}} \\{0,} & {{otherwise}.}\end{matrix} \right.} & {{Eq}\mspace{14mu} (18)}\end{matrix}$

By substituting γ_(ave) from equation (13) and S from equation (14) intoequation (18), the following is obtained:

${\overset{\sim}{\gamma}\left( {j,k} \right)} = \left\{ \begin{matrix}{{\beta \; N_{T}N_{F}},} & {{\gamma \left( {j,k} \right)} \geq {\alpha \; \gamma_{ave}}} \\{0,} & {{otherwise}.}\end{matrix} \right.$

FIG. 2A is a flow diagram of a process 200 to determine the amount oftransmit power to be allocated to each selected transmission channelbased on selective channel inversion, in accordance with an embodimentof the invention. Process 200 may be used if the channel gains H(j,k),the received SNRs γ(j,k), or some other characteristics are availablefor the transmission channels. For clarity, process 200 is describedbelow for the case in which the channel gains are available, and thecase in which the received SNRs are available is shown within brackets.

Initially, the channel gains H(j,k) [or the received SNRs γ(j,k)] of allavailable transmission channels are retrieved, at step 212. A power gainthreshold, αL_(ave), [or an SNR threshold, αγ_(ave)] used to selecttransmission channels for data transmission is also determined, at step214. The threshold may be computed as described in further detail below.

Each available transmission channel is then evaluated for possible use.A (not yet evaluated) available transmission channel is identified forevaluation, at step 216. For the identified transmission channel, adetermination is made whether or not the power gain [or the receivedSNR] for the channel is greater than or equal to the power gainthreshold (i.e., |H(j,k)|²≧αL_(ave)) [or the SNR threshold (i.e.,γ(j,k)≧αγ_(ave)], at step 218. If the identified transmission channelsatisfies the criteria, then it is selected for use, at step 220.Otherwise, if the transmission channel does not satisfy the criteria, itis discarded and not used for data transmission.

A determination is then made whether or not all available transmissionchannels have been evaluated, at step 222. If not, the process returnsto step 216 and another available transmission channel is identified forevaluation. Otherwise, the process proceeds to step 224.

At step 224, a normalization factor {tilde over (b)} [or β] used todistribute the total transmit power among the selected transmissionchannels is determined based on the channel gains [or the received SNRs]of the selected channels, at step 224. This can be achieved as shown inequation (10) [or equation (15)]. A weight {tilde over (W)}(j,k) is nextcomputed for each selected transmission channel, at step 226, based onthe normalization factor and that channel's gain [or SNR]. The weightcan be computed as shown in equation (8) [or equation (16)]. Theweighted transmit power for each selected transmission channel wouldthen be as shown in equation (9) [or equation (17)]. The process thenterminates.

Threshold Selection

The threshold, α, may be selected based on various criteria. In oneembodiment, the threshold is selected to optimize throughput.

Initially, a vector of setpoints (i.e., Z [z₁, z₂, . . . , z_(N)]) and avector of code rates (i.e., R=[r₁, r₂, . . . , r_(N)]) are defined. Eachvector includes N elements corresponding to the number of available coderates, which may be those available for use in the system.Alternatively, N setpoints may be defined based on the operating pointssupported by the system. Each setpoint corresponds to a particularreceived SNR needed to achieve a particular level of performance. In anycase, each code rate r_(n), where 1≦n≦N, corresponds to a respectivesetpoint z_(n), which is the minimum received SNR required for operatingat that code rate for a particular level of performance. The requiredsetpoint, z_(n), may be determined based on computer simulation ormathematical derivation, as is known in the art. The elements in the twovectors R and Z may also be ordered such that {z₁>z₂> . . . >z_(N)} and{r₁>r₂> . . . >r_(N)}.

The channel gains for all available transmission channels are ranked andplaced in a list H(l), where 1≦l≦N_(T)N_(F), such that H(1)=max(|H(j,k)|²), . . . , and H(N_(T)N_(F))=min (|H(j,k)|²).

A sequence {tilde over (b)}(l) of possible normalization factors is alsodefined as follows:

$\begin{matrix}{{{\overset{\sim}{b}()} = \frac{1}{\sum\limits_{i = 1}^{}{{H\left( {j,k} \right)}}^{- 2}}},{1<= \leq {N_{T}{N_{F}.}}}} & {{Eq}\mspace{14mu} (19)}\end{matrix}$

Each element of the sequence {tilde over (b)}(l) may be used as anormalization factor if the l best transmission channels are selectedfor use.

For each code rate r_(n) (where 1≦n≦N), the largest value of l,l_(n,max), is determined such that the received SNR for each of the lbest transmission channels is greater than or equal to the setpointz_(n) corresponding to the code rate r_(n). This condition may beexpressed as:

$\begin{matrix}{{\frac{{\overset{\sim}{b}()}P_{tx}}{\sigma^{2}} \geq z_{n}},} & {{Eq}\mspace{14mu} (20)}\end{matrix}$

where σ² is the received noise power in a single transmission channel.The largest value of l can be identified by evaluating each value of lstarting with 1. For each value of l, the achievable SNR for the l besttransmission channels may be determined as shown by the left argument ofequation (20). This achievable SNR is then compared against the SNR,z_(n), required for that code rate r_(n).

Thus, for each code rate r_(n), each value of l (for l=1, 2, and so on)is evaluated to determine whether the received SNR for each of the lbest transmission channels can achieve the corresponding setpoint z_(n)if the total transmit power is (unevenly) distributed across all lchannels. The largest value of l, l_(n,max), that satisfy this conditionis the most number of transmission channels that may be selected forcode rate r_(n) while achieving the required setpoint z_(n).

The threshold, σ(n), associated with code rate r_(n) may then beexpressed as:

$\begin{matrix}{{\alpha (n)} = {\frac{H\left( _{n,\max} \right)}{L_{ave}}.}} & {{Eq}\mspace{14mu} (21)}\end{matrix}$

The threshold σ(n) optimizes the throughput for code rate r_(n), whichrequires the setpoint z_(n). Since the same code rate is used for allselected transmission channels, the maximum available throughput, T_(n),can be computed as the throughput for each channel (which is r_(n))times the number of selected channels, l_(n,max). The maximum availablethroughput T_(n) for setpoint z_(n) can be expressed as:

T_(n)=l_(n,max)r_(n),  Eq (22)

where the unit for T_(n) is in information bits per modulation symbol.

The optimum throughput for the vector of setpoints can then be given by:

T _(opt)=max(T _(r)).  Eq (23)

As the code rate increases, more information bits may be transmitted permodulation symbol. However, the required SNR also increases, whichrequires more transmit power for each selected transmission channel fora given noise variance. Since the total transmit power is limited, fewertransmission channels may be able to achieve the higher required SNR.Thus, the maximum available throughput for each code rate in the vectormay be computed, and the code rate that provides the highest throughputmay be deemed as the optimum code rate for the specific channelconditions being evaluated. The optimum threshold, α_(opt), is thenequal to the threshold α(n) corresponding to the code rate r_(n) thatresults in T_(opt).

In the above description, the optimum threshold α_(opt) is determinedbased on the channel gains for all transmission channels. If thereceived SNRs are available instead of the channel gains, then thereceived SNRs may be ranked and placed in a list γ(l), where1≦l≦N_(T)N_(F), such that the first element in the list γ(1)=max(γ(j,k)), . . . , and the last element in the list γ(N_(T)N_(R))=min(γ(j,k)). A sequence β(l) may then be determined as:

$\begin{matrix}{{{\beta \; ()} = \frac{1}{\sum\limits_{i = 1}^{}{\gamma (i)}^{- 1}}},} & {{Eq}\mspace{20mu} (24)}\end{matrix}$

For each code rate r_(n) (where 1≦n≦N), the largest value of l,l_(n,max), is determined such that the received SNR for each of the lselected transmission channels is greater than or equal to thecorresponding setpoint z_(n). This condition may be expressed as:

β(l)N _(T) N _(F) ≧z _(n).  Eq (25)

Once the largest value of £, l_(n,max), is determined for code rater_(n), the threshold α(n) associated with this code rate may bedetermined as:

$\begin{matrix}{{\alpha (n)} = {\frac{\gamma \left( _{n,\max} \right)}{\gamma_{ave}}.}} & {{Eq}\mspace{14mu} (26)}\end{matrix}$

The optimum threshold, α_(opt), and the optimum throughput, T_(opt), mayalso be determined as described above.

For the above description, the threshold is selected to optimizethroughput. The threshold may also be selected to optimize otherperformance criteria or metrics, and this is within the scope of theinvention.

FIG. 2B is a flow diagram of a process 250 to determine a threshold αused to select transmission channels for data transmission, inaccordance with an embodiment of the invention. Process 250 may be usedif the channel gains, received SNRs, or some other characteristics areavailable for the transmission channels. For clarity, process 250 isdescribed below for the case in which the channel gains are available,and the case in which the received SNRs are available is shown withinbrackets.

Initially, a vector of setpoints (Z=[z₁, z₂, . . . , z_(N)]) is definedand a vector of code rates (R=[r₁, r₂, . . . , r_(N)]) that supports thecorresponding setpoints is determined, at step 250. The channel gainsH(j,k) [or the received SNRs γ(j,k)] of all available transmissionchannels are retrieved and ranked from the best to the worst, at step252. The sequence {tilde over (b)}(l) [or β(l)] of possiblenormalization factors is then determined based on the channel gains asshown in equation (19) [or based on the received SNRs as shown inequation (24)], at step 254.

Each available code rate is then evaluated via a loop. In the first stepof the loop, a (not yet evaluated) code rate r_(n) is identified forevaluation, at step 256. For the first pass through the loop, theidentified code rate can be the first code rate r₁ in the vector. Forthe identified code rate r_(n), the largest value of l, l_(n,max), isdetermined such that the received SNR for each of the best transmissionchannels is greater than or equal to the corresponding setpoint z_(n),at step 258. This can be performed by computing and satisfying thecondition shown in equation (20) [or equation (25)]. The threshold α(n)associated with setpoint z_(n) is then determined based on the channelgain [or the received SNR] of channel l_(n,max) as shown in equation(21), at step 260. The maximum available throughput, T_(n), for setpointz_(n) can also be determined as shown in equation (22), at step 262.

A determination is then made whether or not all code rates have beenevaluated, at step 264. If not, the process returns to step 256 andanother code rate is identified for evaluation. Otherwise, the optimumthroughput, T_(opt), and the optimum threshold, α_(opt), may bedetermined, at step 266. The process terminates.

Multi-Channel Communication System

FIG. 3 is a diagram of a MIMO communication system 300 capable ofimplementing various aspects and embodiments of the invention. System300 includes a first system 310 (e.g., base station 104 in FIG. 1) incommunication with a second system 350 (e.g., terminal 106). System 300may be operated to employ a combination of antenna, frequency, andtemporal diversity to increase spectral efficiency, improve performance,and enhance flexibility.

At system 310, a data source 312 provides data (i.e., information bits)to a transmit (TX) data processor 314, which (1) encodes the data inaccordance with a particular encoding scheme, (2) interleaves (i.e.,reorders) the encoded data based on a particular interleaving scheme,(3) maps the interleaved bits into modulation symbols for one or moretransmission channels selected for data transmission, and (4)pre-weights the modulation symbols for each selected transmissionchannel. The encoding increases the reliability of the datatransmission. The interleaving provides time diversity for the codedbits, permits the data to be transmitted based on an average SNR for theselected transmission channels, combats fading, and further removescorrelation between coded bits used to form each modulation symbol. Theinterleaving may further provide frequency diversity if the coded bitsare transmitted over multiple frequency subchannels. The pre-weightingeffectively controls the transmit power for each selected transmissionchannel to achieve a desired SNR at the receiver system. In an aspect,the coding, symbol mapping, and pre-weighting may be performed based oncontrol signals provided by a controller 334.

A TX channel processor 320 receives and demultiplexes the weightedmodulation symbols from TX data processor 314 and provides a stream ofweighted modulation symbols for each transmission channel, one weightedmodulation symbol per time slot. TX channel processor 320 may furtherprecondition the weighted modulation symbols for each selectedtransmission channel if full CSI is available.

If OFDM is not employed, TX data processor 314 provides a stream ofweighted modulation symbols for each antenna used for data transmission.And if OFDM is employed, TX data processor 314 provides a stream ofweighted modulation symbol vectors for each antenna used for datatransmission. And if full-CSI processing is performed, TX data processor314 provides a stream of preconditioned modulation symbols orpreconditioned modulation symbol vectors for each antenna used for datatransmission. Each stream is then received and modulated by a respectivemodulator (MOD) 322 and transmitted via an associated antenna 324.

At receiver system 350, a number of receive antennas 352 receive thetransmitted signals and provide the received signals to respectivedemodulators (DEMOD) 354. Each demodulator 354 performs processingcomplementary to that performed at modulator 322. The modulation symbolsfrom all demodulators 354 are provided to a receive (RX) channel/dataprocessor 356 and processed to recover the transmitted data streams. RXchannel/data processor 356 performs processing complementary to thatperformed by TX data processor 314 and TX channel processor 320 andprovides decoded data to a data sink 360. The processing by receiversystem 350 is described in further detail below.

MIMO Transmitter Systems

FIG. 4A is a block diagram of a MIMO transmitter system 310 a, which iscapable of processing data in accordance with an embodiment of theinvention. Transmitter system 310 a is one embodiment of the transmitterportion of system 310 in FIG. 3. System 310 a includes (1) a TX dataprocessor 314 a that receives and processes information bits to provideweighted modulation symbols and (2) a TX channel processor 320 a thatdemultiplexes the modulation symbols for the selected transmissionchannels.

In the embodiment shown in FIG. 4A, TX data processor 314 a includes anencoder 412, a channel interleaver 414, a puncturer 416, a symbolmapping element 418, and a symbol weighting element 420. Encoder 412receives the aggregate information bits to be transmitted and encodesthe received bits in accordance with a particular encoding scheme toprovide coded bits. Channel interleaver 414 interleaves the coded bitsbased on a particular interleaving scheme to provide diversity.Puncturer 416 punctures (i.e., deletes) zero or more of the interleavedcoded bits to provide the desired number of coded bits. Symbol mappingelement 418 maps the unpunctured bits into modulation symbols for theselected transmission channels. And symbol weighting element 420 weighsthe modulation symbols for each selected transmission channel based on arespective weight selected for that channel to provide weightedmodulation symbols.

Pilot data (e.g., data of known pattern) may also be encoded andmultiplexed with the processed information bits. The processed pilotdata may be transmitted (e.g., in a time division multiplexed (TDM)manner) in a subset or all of the selected transmission channels, or ina subset or all of the available transmission channels. The pilot datamay be used at the receiver to perform channel estimation, as describedbelow.

As shown in FIG. 4A, the data encoding, interleaving, and puncturing maybe achieved based on one or more coding control signals, which identifythe specific coding, interleaving, and puncturing schemes to be used.The symbol mapping may be achieved based on a modulation control signalthat identifies the specific modulation scheme to be used. And thesymbol weighting may be achieved based on weights provided for theselected transmission channels.

In one coding and modulation scheme, the coding is achieved by using afixed base code and adjusting the puncturing to achieve the desired coderate, as supported by the SNR of the selected transmission channels. Thebase code may be a Turbo code, a convolutional code, a concatenatedcode, or some other code. The base code may also be of a particular rate(e.g., a rate 1/3 code). For this scheme, the puncturing may beperformed after the channel interleaving to achieve the desired coderate for the selected transmission channels.

Symbol mapping element 416 can be designed to group sets of unpuncturedbits to form non-binary symbols, and to map each non-binary symbol intoa point in a signal constellation corresponding to the modulation schemeselected for the selected transmission channels. The modulation schememay be QPSK, M-PSK, M-QAM, or some other scheme. Each mapped signalpoint corresponds to a modulation symbol.

The encoding, interleaving, puncturing, and symbol mapping attransmitter system 310 a can be performed based on numerous schemes. Onespecific scheme is described in the aforementioned U.S. Pat. No.6,961,388.

The number of information bits that may be transmitted for eachmodulation symbol for a particular level of performance (e.g., onepercent frame error rate or FER) is dependent on the received SNR. Thus,the coding and modulation scheme for the selected transmission channelsmay be determined based on the characteristics of the channels (e.g.,the channel gains, received SNRs, or some other information). Thechannel interleaving may also be adjusted based on the coding controlsignal.

Table 1 lists various combinations of coding rate and modulation schemethat may be used for a number of received SNR ranges. The supported bitrate for each transmission channel may be achieved using any one of anumber of possible combinations of coding rate and modulation scheme.For example, one information bit per modulation symbol may be achievedusing (1) a coding rate of 1/2 and QPSK modulation, (2) a coding rate of1/3 and 8-PSK modulation, (3) a coding rate of 1/4 and 16-QAM, or someother combination of coding rate and modulation scheme. In Table 1,QPSK, 16-QAM, and 64-QAM are used for the listed SNR ranges. Othermodulation schemes such as 8-PSK, 32-QAM, 128-QAM, and so on, may alsobe used and are within the scope of the invention.

TABLE 1 Received SNR # of Information Modulation # of Coded Coding RangeBits/Symbol Symbol Bits/Symbol Rate 1.5-4.4 1 QPSK 2 ½ 4.4-6.4 1.5 QPSK2 ¾  6.4-8.35 2 16-QAM 4 ½ 8.35-10.4 2.5 16-QAM 4 ⅝ 10.4-12.3 3 16-QAM 4¾  12.3-14.15 3.5 64-QAM 6   7/12 14.15-15.55 4 64-QAM 6 ⅔ 15.55-17.354.5 64-QAM 6 ¾ >17.35 5 64-QAM 6 ⅚

The weighted modulation symbols from TX data processor 314 a areprovided to TX channel processor 320 a, which is one embodiment of TXchannel processor 320 in FIG. 3. Within TX channel processor 320 a, ademultiplexer 424 receives and demultiplexes the weighted modulationsymbol into a number of modulation symbol streams, one stream for eachtransmission channel selected to transmit the modulation symbols. Eachmodulation symbol stream is provided to a respective modulator 322. IfOFDM is employed, the weighted modulation symbols at each time slot forall selected frequency subchannels of each transmit antenna are combinedinto a weighted modulation symbol vector. Each modulator 322 convertsthe weighted modulation symbols (for a system without OFDM) or theweighted modulation symbol vectors (for a system with OFDM) into ananalog signal, and further amplifies, filters, quadrature modulates, andupconverts the signal to generate a modulated signal suitable fortransmission over the wireless link.

FIG. 4B is a block diagram of a MIMO transmitter system 310 b, which iscapable of processing data in accordance with another embodiment of theinvention. Transmitter system 310 b is another embodiment of thetransmitter portion of system 310 in FIG. 3. System 310 b includes a TXdata processor 314 b and a TX channel processor 320 b.

In the embodiment shown in FIG. 4B, TX data processor 314 b includesencoder 412, channel interleaver 414, symbol mapping element 418, andsymbol weighting element 420. Encoder 412 receives and encodes theaggregate information bits in accordance with a particular encodingscheme to provide coded bits. The coding may be achieved based on aparticular code and code rate selected by controller 334, as identifiedby the coding control signals. Channel interleaver 414 interleaves thecoded bits, and symbol mapping element 418 maps the interleaved bitsinto modulation symbols for the selected transmission channels. Symbolweighting element 420 weighs the modulation symbols for each selectedtransmission channel based on a respective weight to provide weightedmodulation symbols.

In the embodiment shown in FIG. 4B, transmitter system 310 b is capableof preconditioning the weighted modulation symbols based on full CSI.Within TX channel processor 320 b, a channel MIMO processor 422demultiplexes the weighted modulation symbols into a number of (up toN_(C)) weighted modulation symbol streams, one stream for each spatialsubchannel (i.e., eigenmode) used to transmit the modulation symbols.For full-CSI processing, channel MIMO processor 422 preconditions the(up to N_(c)) weighted modulation symbols at each time slot to generateN_(T) preconditioned modulation symbols, as follows:

$\begin{matrix}{\begin{bmatrix}x_{1} \\x_{2} \\\vdots \\x_{N_{T}}\end{bmatrix} = {\begin{bmatrix}{e_{11},} & {e_{12},} & \ldots & e_{1\; N_{C}} \\{e_{21},} & {e_{22},} & \; & e_{2\; N_{C}} \\\vdots & \; & \ddots & \vdots \\{e_{N_{T}1},} & {e_{N_{T}1},} & \ldots & e_{N_{T}N_{C}}\end{bmatrix} \cdot \begin{bmatrix}b_{1} \\b_{2} \\\vdots \\b_{N_{C}}\end{bmatrix}}} & {{Eq}\mspace{14mu} (27)}\end{matrix}$

where b₁, b₂, . . . and b_(Nc) are respectively the weighted modulationsymbols for the spatial subchannels 1, 2, . . . N_(Nc);e_(ij) are elements of an eigenvector matrix E related to thetransmission characteristics from the transmit antennas to the receiveantennas; andx₁, x₂, . . . x_(N) _(T) are the preconditioned modulation symbols,which can be expressed as:

x ₁ =b ₁ ·e ₁₁ +b ₂ ·e ₁₂ + . . . +b _(N) _(C) ·e _(1N) _(C) ,

x ₂ =b ₁ ·e ₂₁ +b ₂ ·e ₂₂ + . . . +b _(N) _(C) ·e _(2N) _(C) , and

x _(N) _(T) =b ₁ ·e _(N) _(T) ₁ +b ₂ ·e _(N) _(T) ₂ + . . . +b _(N) _(C)·e _(N) _(T) _(N) _(C) .

The eigenvector matrix E may be computed by the transmitter or isprovided to the transmitter by the receiver. The elements of the matrixE are also taken into account in determining the effective channel gainsH(j,k).

For full-CSI processing, each preconditioned modulation symbol, x_(i),for a particular transmit antenna represents a linear combination of theweighted modulation symbols for up to N_(C) spatial subchannels. Foreach time slot, the (up to) N_(T) preconditioned modulation symbolsgenerated by channel MIMO processor 422 are demultiplexed bydemultiplexer 424 and provided to (up to) N_(T) modulators 322. Eachmodulator 322 converts the preconditioned modulation symbols (for asystem without OFDM) or the preconditioned modulation symbol vectors(for a system with OFDM) into a modulated signal suitable fortransmission over the wireless link.

FIG. 4C is a block diagram of a MIMO transmitter system 310 c, whichutilizes OFDM and is capable of processing data in accordance with yetanother embodiment of the invention. Within a TX data processor 314 c,the information bits to be transmitted are demultiplexed by ademultiplexer 428 into a number of (up to N_(L)) frequency subchanneldata streams, one stream for each of the frequency subchannels to beused for the data transmission. Each frequency subchannel data stream isprovided to a respective frequency subchannel data processor 430.

Each data processor 430 processes data for a respective frequencysubchannel of the OFDM system. Each data processor 430 may beimplemented similar to TX data processor 314 a in FIG. 4A, TX dataprocessor 314 b shown in FIG. 4B, or with some other design. In oneembodiment, data processor 430 demultiplexes the frequency subchanneldata stream into a number of data substreams, one data substream foreach spatial subchannel selected for use for the frequency subchannel.Each data substream is then encoded, interleaved, symbol mapped, andweighted to generate modulation symbols for the data substream. Thecoding and modulation for each frequency subchannel data stream or eachdata substream may be adjusted based on the coding and modulationcontrol signals and the weighting may be performed based on the weights.Each data processor 430 thus provides up to N_(C) modulation symbolstreams for up to N_(C) spatial subchannels selected for use for thefrequency subchannel.

For a MIMO system utilizing OFDM, the modulation symbols may betransmitted on multiple frequency subchannels and from multiple transmitantennas. Within a MIMO processor 320 c, the up to N_(C) modulationsymbol streams from each data processor 430 are provided to a respectivesubchannel spatial processor 432, which processes the receivedmodulation symbols based on the channel control and/or the availableCSI. Each spatial processor 432 may simply implement a demultiplexer(such as that shown in FIG. 4A) if full-CSI processing is not performed,or may implement a channel MIMO processor followed by a demultiplexer(such as that shown in FIG. 4B) if full-CSI processing is performed. Fora MIMO system utilizing OFDM, the full-CSI processing (i.e.,preconditioning) may be performed on each frequency subchannel.

Each subchannel spatial processor 432 demultiplexes the up to N_(C)modulation symbols for each time slot into up to N_(T) modulationsymbols for the transmit antennas selected for use for that frequencysubchannel. For each transmit antenna, a combiner 434 receives themodulation symbols for up to N_(L) frequency subchannels selected foruse for that transmit antenna, combines the symbols for each time slotinto a modulation symbol vector V, and provides the modulation symbolvector to the next processing stage (i.e., a respective modulator 322).

MIMO processor 320 c thus receives and processes the modulation symbolsto provide up to N_(T) modulation symbol vectors, V₁ through V_(Nt), onemodulation symbol vector for each transmit antenna selected for use fordata transmission. Each modulation symbol vector V covers a single timeslot, and each element of the modulation symbol vector V is associatedwith a specific frequency subchannel having a unique subcarrier on whichthe modulation symbol is conveyed.

FIG. 4C also shows an embodiment of modulator 322 for OFDM. Themodulation symbol vectors V₁ through V_(Nt) from MIMO processor 320 care provided to modulators 322 a through 322 t, respectively. In theembodiment shown in FIG. 4C, each modulator 322 includes an inverse FastFourier Transform (IFFT) 440, cyclic prefix generator 442, and anupconverter 444.

IFFT 440 converts each received modulation symbol vector into itstime-domain representation (which is referred to as an OFDM symbol)using IFFT. IFFT 440 can be designed to perform the IFFT on any numberof frequency subchannels (e.g., 8, 16, 32, and so on). In an embodiment,for each modulation symbol vector converted to an OFDM symbol, cyclicprefix generator 442 repeats a portion of the time-domain representationof the OFDM symbol to form a “transmission symbol” for a specifictransmit antenna. The cyclic prefix insures that the transmission symbolretains its orthogonal properties in the presence of multipath delayspread, thereby improving performance against deleterious path effects.The implementation of IFFT 440 and cyclic prefix generator 442 is knownin the art and not described in detail herein.

The time-domain representations from each cyclic prefix generator 442(i.e., the transmission symbols for each antenna) are then processed(e.g., converted into an analog signal, modulated, amplified, andfiltered) by upconverter 444 to generate a modulated signal, which isthen transmitted from a respective antenna 324.

OFDM modulation is described in further detail in a paper entitled“Multicarrier Modulation for Data Transmission: An Idea Whose Time HasCome,” by John A. C. Bingham, IEEE Communications Magazine, May 1990,which is incorporated herein by reference.

FIGS. 4A through 4C show three designs of a MIMO transmitter capable ofimplementing various aspects and embodiments of the invention. Theinvention may also be practiced in an OFDM system that does not utilizeMIMO. Numerous other transmitter designs are also capable ofimplementing various inventive techniques described herein, and thesedesigns are also within the scope of the invention. Some of thesetransmitter designs are described in further detail in:

the aforementioned U.S. Pat. No. 6,961,388;

U.S. Patent Application Publication No. 2002-0154705 (now abandoned),entitled “HIGH EFFICIENCY, HIGH PERFORMANCE COMMUNICATIONS SYSTEMEMPLOYING MULTI-CARRIER MODULATION,” filed Mar. 22, 2000;

U.S. Pat. No. 6,771,706, “METHOD AND APPARATUS FOR UTILIZING CHANNELSTATE INFORMATION IN A WIRELESS COMMUNICATION SYSTEM,” filed Mar. 23,2001; and

U.S. Pat. No. 6,785,341, entitled “METHOD AND APPARATUS FOR PROCESSINGDATA IN A MULTIPLE-INPUT MULTIPLE-OUTPUT (MIMO) COMMUNICATION SYSTEMUTILIZING CHANNEL STATE INFORMATION,” filed May 11, 2001.

All of these are assigned to the assignee of the present application andincorporated herein by reference. These patent applications alsodescribe MIMO processing and CSI processing in further detail.

In general, transmitter system 310 codes and modulates data for allselected transmission channels based a particular common coding andmodulation scheme. The modulation symbols are further weighted byweights assigned to the selected transmission channels such that thedesired level of performance is achieved at the receiver. The techniquesdescribed herein are applicable for multiple parallel transmissionchannels supported by MIMO, OFDM, or any other communication scheme(e.g., a CDMA scheme) capable of supporting multiple paralleltransmission channels.

MIMO Receiver Systems

FIG. 5 is a block diagram of a MIMO receiver system 350 a capable ofreceiving data in accordance with an embodiment of the invention.Receiver system 350 a is one specific embodiment of receiver system 350in FIG. 3 and implements the successive cancellation receiver processingtechnique to receive and recover the transmitted signals. Thetransmitted signals from (up to) N_(T) transmit antennas are received byeach of N_(R) antennas 352 a through 352 r and routed to a respectivedemodulator (DEMOD) 354 (which is also referred to as a front-endprocessor).

Each demodulator 354 conditions (e.g., filters and amplifies) arespective received signal, downconverts the conditioned signal to anintermediate frequency or baseband, and digitizes the downconvertedsignal to provide samples. Each demodulator 354 may further demodulatethe samples with a received pilot to generate a stream of receivedmodulation symbols, which is provided to an RX channel/data processor356 a.

If OFDM is employed for the data transmission, each demodulator 354further performs processing complementary to that performed by modulator322 shown in FIG. 4C. In this case, each demodulator 354 includes an FFTprocessor (not shown) that generates transformed representations of thesamples and provides a stream of modulation symbol vectors. Each vectorincludes up to N_(L) modulation symbols for up to N_(L) frequencysubchannels selected for use, and one vector is provided for each timeslot. The modulation symbol vector streams from the FFT processors ofall N_(R) demodulators are then provided to a demultiplexer (not shownin FIG. 5), which “channelizes” the modulation symbol vector stream fromeach FFT processor into up to N_(L) modulation symbol streamscorresponding to the number of frequency subchannels used for the datatransmission. For the transmit processing scheme in which each frequencysubchannel is independently processed (e.g., as shown in FIG. 4C), thedemultiplexer further provides each of up to N_(L) modulation symbolstreams to a respective RX MIMO/data processor 356 a.

For a MIMO system utilizing OFDM, one RX MIMO/data processor 356 a maybe used to process the set of N_(R) modulation symbol streams from theN_(R) received antennas for each of up to N_(L) frequency subchannelsused for data transmission. Alternatively, the set of modulation symbolstreams associated with each frequency subchannel may be processedseparately by a single RX channel/data processor 356 a. And for a MIMOsystem not utilizing OFDM, one RX MIMO/data processor 356 a may be usedto process the N_(R) modulation symbol streams from the N_(R) receivedantennas.

In the embodiment shown in FIG. 5, RX channel/data processor 356 a(which is one embodiment of RX channel/data processor 356 in FIG. 3)includes a number of successive (i.e., cascaded) receiver processingstages 510, one stage for each of the transmitted data streams to berecovered by receiver system 350 a. In one transmit processing scheme,one data stream is transmitted on each transmission channel used fordata transmission to receiver system 350 a, and each data stream isindependently processed (e.g., with its own coding and modulationscheme) and transmitted from a respective transmit antenna. For thistransmit processing scheme, the number of data streams to be recoveredfor each OFDM subchannel by receiver system 350 a is equal to the numberof transmission channels, which is also equal to the number of transmitantennas used for data transmission to receiver system 350 a (which maybe a subset of the available transmit antennas). For clarity, RXchannel/data processor 356 a is described for this transmit processingscheme.

Each receiver processing stage 510 (except for the last stage 510 n)includes a channel MIMO/data processor 520 coupled to an interferencecanceller 530, and the last stage 510 n includes only channel MIMO/dataprocessor 520 n. For the first receiver processing stage 510 a, channelMIMO/data processor 520 a receives and processes the N_(R) modulationsymbol streams from demodulators 354 a through 354 r to provide adecoded data stream for the first transmission channel (or the firsttransmitted signal). And for each of the second through last stages 510b through 510 n, channel MIMO/data processor 520 for that stage receivesand processes the N_(R) modified symbol streams from the interferencecanceller 520 in the preceding stage to derive a decoded data stream forthe transmission channel being processed by that stage. Each channelMIMO/data processor 520 further provides CSI (e.g., the SNR) for theassociated transmission channel.

For the first receiver processing stage 510 a, interference canceller530 a receives the N_(R) modulation symbol streams from all N_(R)demodulators 354. And for each of the second through second-to-laststages, interference canceller 530 receives the N_(R) modified symbolstreams from the interference canceller in the preceding stage. Eachinterference canceller 530 also receives the decoded data stream fromchannel MIMO/data processor 520 within the same stage, and performs theprocessing (e.g., coding, interleaving, modulation, channel response,and so on) to derive N_(R) remodulated symbol streams that are estimatesof the interference components of the received modulation symbol streamsdue to this decoded data stream. The remodulated symbol streams are thensubtracted from the received modulation symbol streams to derive N_(R)modified symbol streams that include all but the subtracted (i.e.,canceled) interference components. The N_(R) modified symbol streams arethen provided to the next stage.

In FIG. 5, a controller 540 is shown coupled to RX channel/dataprocessor 356 a and may be used to direct various steps in thesuccessive cancellation receiver processing performed by processor 356a.

FIG. 5 shows a receiver structure that may be used in a straightforwardmanner when each data stream is transmitted over a respective transmitantenna (i.e., one data stream corresponding to each transmittedsignal). In this case, each receiver processing stage 510 may beoperated to recover one of the transmitted signals targeted for receiversystem 350 a and provide the decoded data stream corresponding to therecovered transmitted signal.

For some other transmit processing schemes, a data stream may betransmitted over multiple transmit antennas, frequency subchannels,and/or time intervals to provide spatial, frequency, and time diversity,respectively. For these schemes, the receiver processing initiallyderives a received modulation symbol stream for the transmitted signalon each transmit antenna of each frequency subchannel. Modulationsymbols for multiple transmit antennas, frequency subchannels, and/ortime intervals may then be combined in a complementary manner as thedemultiplexing performed at the transmitter system. The stream ofcombined modulation symbols is then processed to provide thecorresponding decoded data stream.

FIG. 6A is a block diagram of an embodiment of channel MIMO/dataprocessor 520 x, which is one embodiment of channel MIMO/data processor520 in FIG. 5. In this embodiment, channel MIMO/data processor 520 xincludes a spatial/space-time processor 610, a CSI processor 612, aselector 614, a demodulation element 618, a de-interleaver 618, and adecoder 620.

Spatial/space-time processor 610 performs linear spatial processing onthe N_(R) received signals for a non-dispersive MIMO channel (i.e., withflat fading) or space-time processing on the N_(R) received signals fora dispersive MIMO channel (i.e., with frequency selective fading). Thespatial processing may be achieved using linear spatial processingtechniques such as a channel correlation matrix inversion (CCMI)technique, a minimum mean square error (MMSE) technique, and others.These techniques may be used to null out the undesired signals or tomaximize the received SNR of each of the constituent signals in thepresence of noise and interference from the other signals. Thespace-time processing may be achieved using linear space-time processingtechniques such as a MMSE linear equalizer (MMSE-LE), a decisionfeedback equalizer (DFE), a maximum-likelihood sequence estimator(MLSE), and others. The CCMI, MMSE, MMSE-LE, and DFE techniques aredescribed in further detail in the aforementioned U.S. Pat. No.6,785,341. The DFE and MLSE techniques are also described in furtherdetail by S. L. Ariyavistakul et al. in a paper entitled “OptimumSpace-Time Processors with Dispersive Interference: Unified Analysis andRequired Filter Span,” IEEE Trans. on Communication, Vol. 7, No. 7, July1999, and incorporated herein by reference.

CSI processor 612 determines the CSI for each of the transmissionchannels used for data transmission. For example, CSI processor 612 mayestimate a noise covariance matrix based on the received pilot signalsand then compute the SNR of the k-th transmission channel used for thedata stream to be decoded. The SNR can be estimated similar toconventional pilot assisted single and multi-carrier systems, as isknown in the art. The SNR for all of the transmission channels used fordata transmission may comprise the CSI that is reported back to thetransmitter system for this transmission channel. CSI processor 612 mayfurther provide to selector 614 a control signal that identifies theparticular data stream to be recovered by this receiver processingstage.

Selector 614 receives a number of symbol streams from spatial/space-timeprocessor 610 and extracts the symbol stream corresponding to the datastream to be decoded, as indicated by the control signal from CSIprocessor 612. The extracted stream of modulation symbols is thenprovided to a demodulation element 614.

For the embodiment shown in FIG. 6 in which the data stream for eachtransmission channel is independently coded and modulated based on thechannel's SNR, the recovered modulation symbols for the selectedtransmission channel are demodulated in accordance with a demodulationscheme (e.g., M-PSK, M-QAM) that is complementary to the modulationscheme used for the transmission channel. The demodulated data fromdemodulation element 616 is then de-interleaved by a de-interleaver 618in a complementary manner to that performed by channel interleaver 614,and the de-interleaved data is further decoded by a decoder 620 in acomplementary manner to that performed by encoder 612. For example, aTurbo decoder or a Viterbi decoder may be used for decoder 620 if Turboor convolutional coding, respectively, is performed at the transmittersystem. The decoded data stream from decoder 620 represents an estimateof the transmitted data stream being recovered.

FIG. 6B is a block diagram of an interference canceller 530 x, which isone embodiment of interference canceller 530 in FIG. 5. Withininterference canceller 530 x, the decoded data stream from the channelMIMO/data processor 520 within the same stage is re-encoded,interleaved, and re-modulated by a channel data processor 628 to provideremodulated symbols, which are estimates of the modulation symbols atthe transmitter system prior to the MIMO processing and channeldistortion. Channel data processor 628 performs the same processing(e.g., encoding, interleaving, and modulation) as that performed at thetransmitter system for the data stream. The remodulated symbols are thenprovided to a channel simulator 630, which processes the symbols withthe estimated channel response to provide estimates, î ^(k), of theinterference due the decoded data stream. The channel response estimatemay be derived based on the pilot and/or data transmitted by thetransmitter system and in accordance with the techniques described inthe aforementioned U.S. Pat. No. 6,785,341.

The N_(R) elements in the interference vector î ^(k) correspond to thecomponent of the received signal at each of the N_(R) receive antennasdue to symbol stream transmitted on the k-th transmit antenna. Eachelement of the vector represents an estimated component due to thedecoded data stream in the corresponding received modulation symbolstream. These components are interference to the remaining (not yetdetected) transmitted signals in the N_(R) received modulation symbolstreams (i.e., the vector r ^(k)), and are subtracted (i.e., canceled)from the received signal vector r ^(k) by a summer 632 to provide amodified vector r ^(k+1) having the components from the decoded datastream removed. The modified vector r ^(k+1) is provided as the inputvector to the next receiver processing stage, as shown in FIG. 5.

Various aspects of the successive cancellation receiver processing aredescribed in further detail in the aforementioned U.S. Pat. No.6,785,341.

FIG. 7 is a block diagram of a MIMO receiver system 350 b capable ofreceiving data in accordance with another embodiment of the invention.The transmitted signals from (up to) N_(T) transmit antennas arereceived by each of N_(R) antennas 352 a through 352 r and routed to arespective demodulator 354. Each demodulator 354 conditions, processes,and digitizes a respective received signal to provide samples, which areprovided to a RX MIMO/data processor 356 b.

Within RX MIMO/data processor 356 b, the samples for each receiveantenna are provided to a respective FFT processor 710, which generatestransformed representations of the received samples and provides arespective stream of modulation symbol vectors. The streams ofmodulation symbol vector from FFT processors 710 a through 710 r arethen provided to a processor 720. Processor 720 channelizes the streamof modulation symbol vectors from each FFT processor 710 into a numberof up to N_(L) subchannel symbol streams. Processor 720 may furtherperform spatial processing or space-time processing on the subchannelsymbol streams to provide post-processed modulation symbols.

For each data stream transmitted over multiple frequency subchannelsand/or multiple spatial subchannels, processor 720 further combines themodulation symbols for all frequency and spatial subchannels used forthe transmission of the data stream into one post-processed modulationsymbol stream, which is then provided to a data stream processor 730.Each data stream processor 730 performs demodulation, deinterleaving,and decoding complementary to that performed on the data stream at thetransmitter unit and provides a respective decoded data stream.

Receiver systems that employ the successive cancellation receiverprocessing technique and those that do not employ the successivecancellation receiver processing technique may be used to receive,process, and recover the transmitted data streams. Some receiver systemscapable of processing signals received over multiple transmissionchannels are described in:

the aforementioned U.S. Pat. No. 6,961,388;

the aforementioned U.S. Pat. No. 6,771,706; and

U.S. Patent Application Publication No. 2002-0154705 (now abandoned),entitled “HIGH EFFICIENCY, HIGH PERFORMANCE COMMUNICATIONS SYSTEMEMPLOYING MULTI-CARRIER MODULATION,” filed Mar. 22, 2000, All of theseare assigned to the assignee of the present invention and incorporatedherein by reference.

Obtaining CSI for the Transmitter System

For simplicity, various aspects and embodiments of the invention havebeen described wherein the CSI comprises SNR. In general, the CSI maycomprise any type of information that is indicative of thecharacteristics of the communication link. Various types of informationmay be provided as CSI, some examples of which are described below.

In one embodiment, the CSI comprises signal-to-noise-plus-interferenceratio (SNR), which is derived as the ratio of the signal power over thenoise plus interference power. The SNR is typically estimated andprovided for each transmission channel used for data transmission (e.g.,each transmit data stream), although an aggregate SNR may also beprovided for a number of transmission channels. The SNR estimate may bequantized to a value having a particular number of bits. In oneembodiment, the SNR estimate is mapped to an SNR index, e.g., using alook-up table.

In another embodiment, the CSI comprises power control information foreach spatial subchannel of each frequency subchannel. The power controlinformation may include a single bit for each transmission channel toindicate a request for either more power or less power, or it mayinclude multiple bits to indicate the magnitude of the change of powerlevel requested. In this embodiment, the transmitter system may make useof the power control information fed back from the receiver systems todetermine which transmission channels to select, and what power to usefor each transmission channel.

In yet another embodiment, the CSI comprises signal power andinterference plus noise power. These two components may be separatelyderived and provided for each transmission channel used for datatransmission.

In yet another embodiment, the CSI comprises signal power, interferencepower, and noise power. These three components may be derived andprovided for each transmission channel used for data transmission.

In yet another embodiment, the CSI comprises signal-to-noise ratio plusa list of interference powers for each observable interference term.This information may be derived and provided for each transmissionchannel used for data transmission.

In yet another embodiment, the CSI comprises signal components in amatrix form (e.g., N_(T)×N_(R) complex entries for all transmit-receiveantenna pairs) and the noise plus interference components in matrix form(e.g., N_(T)×N_(R) complex entries). The transmitter system may thenproperly combine the signal components and the noise plus interferencecomponents for the appropriate transmit-receive antenna pairs to derivethe quality for each transmission channel used for data transmission(e.g., the post-processed SNR for each transmitted data stream, asreceived at the receiver systems).

In yet another embodiment, the CSI comprises a data rate indicator foreach transmit data stream. The quality of a transmission channel to beused for data transmission may be determined initially (e.g., based onthe SNR estimated for the transmission channel) and a data ratecorresponding to the determined channel quality may then be identified(e.g., based on a look-up table). The identified data rate is indicativeof the maximum data rate that may be transmitted on the transmissionchannel for the required level of performance. The data rate is thenmapped to and represented by a data rate indicator (DRI), which can beefficiently coded. For example, if (up to) seven possible data rates aresupported by the transmitter system for each transmit antenna, then a3-bit value may be used to represent the DRI where, e.g., a zero mayindicate a data rate of zero (i.e., don't use the transmit antenna) and1 through 7 may be used to indicate seven different data rates. In atypical implementation, the quality measurements (e.g., SNR estimates)are mapped directly to the DRI based on, e.g., a look-up table.

In yet another embodiment, the CSI comprises an indication of theparticular processing scheme to be used at the transmitter system foreach transmit data stream. In this embodiment, the indicator mayidentify the particular coding scheme and the particular modulationscheme to be used for the transmit data stream such that the desiredlevel of performance is achieved.

In yet another embodiment, the CSI comprises a differential indicatorfor a particular measure of quality for a transmission channel.Initially, the SNR or DRI or some other quality measurement for thetransmission channel is determined and reported as a referencemeasurement value. Thereafter, monitoring of the quality of thetransmission channel continues, and the difference between the lastreported measurement and the current measurement is determined. Thedifference may then be quantized to one or more bits, and the quantizeddifference is mapped to and represented by the differential indicator,which is then reported. The differential indicator may indicate toincrease or decrease the last reported measurement by a particular stepsize (or to maintain the last reported measurement). For example, thedifferential indicator may indicate that (1) the observed SNR for aparticular transmission channel has increased or decreased by aparticular step size, or (2) the data rate should be adjusted by aparticular amount, or some other change. The reference measurement maybe transmitted periodically to ensure that errors in the differentialindicators and/or erroneous reception of these indicators do notaccumulate.

In yet another embodiment, the CSI comprise channel gain for eachavailable transmission channel as estimated at the receiver system basedon signals transmitted by the transmitter system.

Other forms of CSI may also be used and are within the scope of theinvention. In general, the CSI includes sufficient information inwhatever form that may be used to (1) select a set of transmissionchannels that will result in optimum or near optimum throughput, (2)determine a weighting factor for each selected transmission channel thatresults in equal or near equal received SNRs, and (3) infer an optimumor near optimum code rate for each selected transmission channel.

The CSI may be derived based on the signals transmitted from thetransmitter system and received at the receiver systems. In anembodiment, the CSI is derived based on a pilot reference included inthe transmitted signals. Alternatively or additionally, the CSI may bederived based on the data included in the transmitted signals. Althoughdata may be transmitted on only the selected transmission channels,pilot data may be transmitted on unselected transmission channels toallow the receiver systems to estimate the channel characteristics.

In yet another embodiment, the CSI comprises one or more signalstransmitted from the receiver systems to the transmitter system. In somesystems, a degree of correlation may exist between the uplink anddownlink (e.g. time division duplexed (TDD) systems where the uplink anddownlink share the same band in a time division multiplexed manner). Inthese systems, the quality of the uplink may be estimated (to arequisite degree of accuracy) based on the quality of the downlink, andvice versa, which may be estimated based on signals (e.g., pilotsignals) transmitted from the receiver systems. The pilot signals wouldthen represent a means for which the transmitter system could estimatethe CSI as observed at the receiver systems. For this type of CSI, noreporting of channel characteristics is necessary.

The signal quality may be estimated at the transmitter system based onvarious techniques. Some of these techniques are described in thefollowing patents, which are assigned to the assignee of the presentapplication and incorporated herein by reference:

U.S. Pat. No. 5,799,005, entitled “SYSTEM AND METHOD FOR DETERMININGRECEIVED PILOT POWER AND PATH LOSS IN A CDMA COMMUNICATION SYSTEM,”issued Aug. 25, 1998,

U.S. Pat. No. 5,903,554, entitled “METHOD AND APPARATUS FOR MEASURINGLINK QUALITY IN A SPREAD SPECTRUM COMMUNICATION SYSTEM,” issued May 11,1999,

U.S. Pat. Nos. 5,056,109, and 5,265,119, both entitled “METHOD ANDAPPARATUS FOR CONTROLLING TRANSMISSION POWER IN A CDMA CELLULAR MOBILETELEPHONE SYSTEM,” respectively issued Oct. 8, 1991 and Nov. 23, 1993,and

U.S. Pat. No. 6,097,972, entitled “METHOD AND APPARATUS FOR PROCESSINGPOWER CONTROL SIGNALS IN CDMA MOBILE TELEPHONE SYSTEM,” issued Aug. 1,2000.

Methods for estimating a single transmission channel based on a pilotsignal or a data transmission may also be found in a number of papersavailable in the art. One such channel estimation method is described byF. Ling in a paper entitled “Optimal Reception, Performance Bound, andCutoff-Rate Analysis of References-Assisted Coherent CDMA Communicationswith Applications,” IEEE Transaction On Communication, October 1999.

Various types of information for CSI and various CSI reportingmechanisms are also described in:

U.S. Pat. No. 6,574,211, entitled “METHOD AND APPARATUS FOR HIGH RATEPACKET DATA TRANSMISSION,” filed Nov. 3, 1997, assigned to the assigneeof the present application; and

“TIE/EIA/IS-856 cdma2000 High Rate Packet Data Air InterfaceSpecification”, both of which are incorporated herein by reference.

The CSI may be reported back to the transmitter using various CSItransmission schemes. For example, the CSI may be sent in full,differentially, or a combination thereof. In one embodiment, CSI isreported periodically, and differential updates are sent based on theprior transmitted CSI. In another embodiment, the CSI is sent only whenthere is a change (e.g., if the change exceeds a particular threshold),which may lower the effective rate of the feedback channel. As anexample, the SNRs may be sent back (e.g., differentially) only when theychange. For an OFDM system (with or without MIMO), correlation in thefrequency domain may be exploited to permit reduction in the amount ofCSI to be fed back. As an example for an OFDM system, if the SNRcorresponding to a particular spatial subchannel for M frequencysubchannels is the same, the SNR and the first and last frequencysubchannels for which this condition is true may be reported. Othercompression and feedback channel error recovery techniques to reduce theamount of data to be fed back for CSI may also be used and are withinthe scope of the invention.

Referring back to FIG. 3, the CSI (e.g., the received SNR) determined byRX channel/data processor 356 is provided to a TX data processor 362,which processes the CSI and provides processed data to one or moremodulators 354. Modulators 354 further condition the processed data andtransmit the CSI back to transmitter system 310 via a reverse channel.

At system 310, the transmitted feedback signal is received by antennas324, demodulated by demodulators 322, and provided to a RX dataprocessor 332. RX data processor 332 performs processing complementaryto that performed by TX data processor 362 and recovers the reportedCSI, which is then provided to controller 334.

Controller 334 uses the reported CSI to perform a number of functionsincluding (1) selecting the set of N_(S) best available transmissionchannels for data transmission, (2) determining the coding andmodulation scheme to be used for data transmission on the selectedtransmission channels, and (3) determining the weights to be used forthe selected transmission channels. Controller 334 may select thetransmission channels to achieve high throughput or based on some otherperformance criteria or metrics, and may further determine the thresholdused to select the transmission channels, as described above.

The characteristics (e.g., channel gains or received SNRs) of thetransmission channels available for data transmission may be determinedbased on various techniques as described above and provided to thetransmitter system. The transmitter system may then use the informationto select the set of N_(S) best transmission channels, properly code andmodulate the data, and further weight the modulation symbols.

The techniques described herein may be used for data transmission on thedownlink from a base station to one or more terminals, and may also beused for data transmission on the uplink from each of one or moreterminals to a base station. For the downlink, transmitter system 310 inFIGS. 3, 4A, and 4B may represent part of a base station and receiversystem 350 in FIGS. 3, 5, and 6 may represent part of a terminal And forthe uplink, transmitter system 310 in FIGS. 3, 4A, and 4B may representpart of a terminal and receiver system 350 in FIGS. 3, 5, and 6 mayrepresent part of a base station.

The elements of the transmitter and receiver systems may be implementedwith one or more digital signal processors (DSP), application specificintegrated circuits (ASIC), processors, microprocessors, controllers,microcontrollers, field programmable gate arrays (FPGA), programmablelogic devices, other electronic units, or any combination thereof. Someof the functions and processing described herein may also be implementedwith software executed on a processor. Certain aspects of the inventionmay also be implemented with a combination of software and hardware. Forexample, computations to determine the threshold, α, and to selecttransmission channels may be performed based on program codes executedon a processor (controller 334 in FIG. 3).

Headings are included herein for reference and to aid in the locatingcertain sections. These heading are not intended to limit the scope ofthe concepts described therein under, and these concepts may haveapplicability in other sections throughout the entire specification.

The previous description of the disclosed embodiments is provided toenable any person skilled in the art to make or use the presentinvention. Various modifications to these embodiments will be readilyapparent to those skilled in the art, and the generic principles definedherein may be applied to other embodiments without departing from thespirit or scope of the invention. Thus, the present invention is notintended to be limited to the embodiments shown herein but is to beaccorded the widest scope consistent with the principles and novelfeatures disclosed herein.

1. A method for transmitting data by a transmitter over multipletransmission channels to a receiver in a multi-channel communicationsystem, comprising: selecting one or more available transmissionchannels from a plurality of transmission channels for use for datatransmission, wherein the selected transmissions channels have signal tonoise ratios (SNRs) at or above a threshold value; coding and modulatingdata for transmission over the selected transmission channels using acommon coding and modulation scheme to provide modulation symbols;weighting the modulation symbols for each selected transmission channelbased on channel state information (CSI) for the selected channels, inan effort to distribute the total available transmit power across theselected transmission channels to achieve similar received signalquality; and transmitting the weighted modulation symbols on theselected transmission channels.
 2. The method of claim 1, wherein themulti-channel communication system is a multiple-input multiple-output(MIMO) that utilizes orthogonal frequency division modulation (OFDM). 3.The method of claim 1, wherein the available transmission channels areselected from one of a group of transmission channels and each groupcorresponds to a respective transmit antenna.
 4. The method of claim 1,wherein the available transmission channels are selected from one of agroup of transmission channels and each group corresponds to a pluralityof frequency subchannels for a corresponding transmit antenna.
 5. Themethod of claim 1, further comprising deriving the weight for eachselected transmission channel based on a normalization factor, which isdetermined based on the characteristics of the selected transmissionchannels.
 6. A transmitter for use in a receiver in a multi-channelcommunication system, comprising: a controller configured to select oneor more available transmission channels from a plurality of transmissionchannels for use for data transmission, wherein the selectedtransmissions channels have signal to noise ratios (SNRs) at or above athreshold value; a transmit data processor configured to code andmodulate data for transmission over the selected transmission channelsusing a common coding and modulation scheme to provide modulationsymbols, weight the modulation symbols for each selected transmissionchannel based on channel state information (CSI) for the selectedchannels, in an effort to distribute the total available transmit poweracross the selected transmission channels to achieve similar receivedsignal quality, and transmit the weighted modulation symbols on theselected transmission channels.
 7. The transmitter of claim 6, whereinthe multi-channel communication system is a multiple-inputmultiple-output (MIMO) that utilizes orthogonal frequency divisionmodulation (OFDM).
 8. The transmitter of claim 6, wherein the availabletransmission channels are selected from one of a group of transmissionchannels and each group corresponds to a respective transmit antenna. 9.The transmitter of claim 6, wherein the available transmission channelsare selected from one of a group of transmission channels and each groupcorresponds to a plurality of frequency subchannels for a correspondingtransmit antenna.
 10. The transmitter of claim 1, further comprisinglogic configured to derive the weight for each selected transmissionchannel based on a normalization factor, which is determined based onthe characteristics of the selected transmission channels.
 11. Anapparatus for transmitting data by a transmitter over multipletransmission channels to a receiver in a multi-channel communicationsystem, comprising: means for selecting one or more availabletransmission channels from a plurality of transmission channels for usefor data transmission, wherein the selected transmissions channels havesignal to noise ratios (SNRs) at or above a threshold value; means forcoding and modulating data for transmission over the selectedtransmission channels using a common coding and modulation scheme toprovide modulation symbols; means for weighting the modulation symbolsfor each selected transmission channel based on channel stateinformation (CSI) for the selected channels, in an effort to distributethe total available transmit power across the selected transmissionchannels to achieve similar received signal quality; and means fortransmitting the weighted modulation symbols on the selectedtransmission channels.
 12. The apparatus of claim 1, wherein themulti-channel communication system is a multiple-input multiple-output(MIMO) that utilizes orthogonal frequency division modulation (OFDM).13. The apparatus of claim 1, wherein the available transmissionchannels are selected from one of a group of transmission channels andeach group corresponds to a respective transmit antenna.
 14. Theapparatus of claim 1, wherein the available transmission channels areselected from one of a group of transmission channels and each groupcorresponds to a plurality of frequency subchannels for a correspondingtransmit antenna.
 15. The apparatus of claim 11, further comprisingmeans for deriving the weight for each selected transmission channelbased on a normalization factor, which is determined based on thecharacteristics of the selected transmission channels.